High probability capture of slow trigger rate acquisitions in a digital oscilloscope with variable intensity rasterization

ABSTRACT

A new oscilloscope design improves the processing of acquired voltage-versus-time data through the efficient high speed acquisition and rasterization of such data into a form that includes multiple-bits-per-pixel intensity information. The multi-bit-per-pixel variable intensity rasterizer is optimized for maximum throughput and most efficient use of memory bandwidth. In the presence of faltering trigger rates, rasterization interruption provides a high probability of capturing the data associated with the slow triggers. Circuitry is provided to compensate for acquisition time and amplitude non-linearities. Many-bits-per-pixel intensity information is mapped into a fewer-bits-per-pixel format by a controllable transfer function that provides multiple viewing capabilities for the operator. Another mode of operation emphasizes infrequent events over commonly occurring ones using variations in brightness or color.

FIELD OF THE INVENTION

This invention relates to the processing of acquired voltage-versus-timedata representative of the activity of a signal under observation into aform that is suitable for display by a digital oscilloscope, and moreparticularly to the efficient high speed acquisition and rasterizationof such data into a form that includes multiple-bits-per-pixel intensityinformation for variable intensity displays and the capability ofproviding a high probability of capturing slow trigger rate acquisitionsby rasterization interruption at appropriate times.

Statement Regarding Federally Sponsored Research and Development [NotApplicable] BACKGROUND OF THE INVENTION

Digital oscilloscopes generally use raster scan displays to present theactivity of electrical signals to their users. Each raster scan display,such as those seen every day on computer screens, consists of a twodimensional array of pixels, with each pixel location being uniquelydefined by a row number and column number. The simplest and lowest costversions of such displays are “single bit” displays, in that the memoryfrom which they derive the information to be displayed only has one bitof intensity information associated with each pixel. In such a displaythat single bit of information determines whether the pixel associatedwith it is either “on” or “off”, with “on” dictating that apredetermined amount of intensity is to be used to illuminate the pixeland “off” indicating that the pixel is not to be illuminated at all.

The more complex and expensive alternative to a single bit display is amulti-bit display, which can provide variable intensity (also known as“gray-scale”) or color variations as a substitute indicator ofbrightness. The memory locations associated with each pixel of avariable intensity display contain multiple bits of intensityinformation, indicating the number of varying intensity levels withwhich they can be illuminated. Like the pixels of single bit displays,those of multi-bit displays can exhibit an “off” or dark state, butinstead of one value of illumination, they have multiple values.Typically, the number of values available is 2^(N−)1, where N is thememory depth at each address of the raster memory. Thus, for example, afour bit deep raster scan memory can support fifteen levels of partialthrough maximum illumination, as well as the dark or “off” state. Pixelintensity can also be translated into differing colors, as well asintensity or “brightness”.

With this larger amount of data, multi-bit displays can convey moreinformation about the behavior of electrical signal waveforms underobservation, particularly if the signal is not perfectly repetitive andtherefore has less activity in some portions than others. U.S. Pat. No.4,940,931 to Katayama et al. for “Digital Waveform Measuring ApparatusHaving A Shading-tone Display”, hereby incorporated by reference,describes a system for producing digital variable intensity displays.

Typically, digital oscilloscopes acquire information about the behaviorof a circuit node by periodically sampling the voltage present at thenode. The oscilloscope probe tip is placed in contact with the node andthe probe and front end of the oscilloscope precisely replicate thesignal, or some predetermined fraction or multiple of the signal, andpresent it to an analog-to-digital converter. The output of theanalog-to-digital converter is a series of multi-bit digital words thatare stored in an acquisition memory. Successively acquired samples arestored at sequentially related addresses in the acquisition memory, andare thereby related to a time scale. Those addresses will eventually beconverted back to a time scale, one of which is represented ashorizontal distance along the x-axis of the oscilloscope's raster scandisplay.

In a typical digital oscilloscope, voltage amplitude values derived fromthe data contents of an acquisition memory location determine thevertical location (row number) of an illuminated pixel, while timevalues derived from the addresses of the acquisition memory determinethe horizontal location (column number). The process of expanding thecontents and addresses of an acquisition memory to produce contents fora two dimensional raster memory is known as “rasterization”.

The output of a rasterization process is usually combined with somepreexisting content of a raster memory, and the resulting compositeraster contents may thereafter be regularly subjected to some sort ofdecay process. For more information about digital persistence and decay,refer to the following U.S. Pat. Nos., hereby incorporated by reference:5,440,676 to Alappat et al. for “Raster Scan Waveform Display RasterizerWith Pixel Intensity Gradation”; 5,387,896 to Alappat et al. for“Rasterscan Display With Adaptive Decay”; 5,254,983 to Long et al. for“Digitally Synthesized Gray Scale For Raster Scan OscilloscopeDisplays”.

For any particular combination of settings of the oscilloscope displayand acquired waveform data, there will be some function that maps theacquired data points into the time (x-axis) versus voltage (y-axis)display raster. That mapping function will include some ratio betweenthe number of samples to be mapped and the number of pixel columns inthe raster display. While that ratio can be 1:1, it will usually be N:1or 1:N If there are more data points than columns of pixels into whichthey must be mapped, some form of data compression and/or decimationwill be utilized. Decimation means discarding all but every Nth datapoint, thereby forgoing part of the available information. Compression,on the other hand, means mapping data from multiple time locations inthe acquisition memory into one horizontal location, i.e., a singlecolumn of pixels, in the raster scan display. If there are fewer datapoints than columns of pixels, i.e., the 1:N case above, then some sortof interpolation or equivalent time sampling is generally used. In thecase of the present inventions, equivalent time sampling, which will bediscussed in much greater detail below, is used.

For many years, digital oscilloscopes were limited in the percentage ofactivity at the probe tip that they effectively processed and displayedfor the user. Although less sophisticated users and those who were onlyfamiliar with analog oscilloscopes had the impression that they wereseeing most or all of the activity at their digital oscilloscope's probetip, in many circumstances the display was really only showing a smallpercentage of the actual activity occurring there. This was becausethese oscilloscopes spent a lot more time processing signals than theydid acquiring them. If a signal is perfectly repetitive, this loss of“live time” is not a problem, since one waveform looks just likeanother. However, when a signal is displaying some sort of intermittentanomalous behavior, a low percentage of live time may prevent such ananomaly from being detected. Therefore, increasing the waveformthroughput and the percentage of signal activity at the probe tip thatcan actually be observed by the user has increasingly been the goal ofthe more recent digital oscilloscope designs. U.S. Pat. No. 5,412,579 toMeadows, et al. for “Slow Display Method for Digital Oscilloscope WithFast Acquisition System”, hereby incorporated by reference, describes anoscilloscope system in which acquisitions are composited intoalternating (also known as “ping-ponging”) display buffers, so thatwhile the contents of one display buffer is being used as the source ofdata being displayed, the other one is being used to gather andcomposite more data. However, the slow display of the design describedin this patent only provided a single bit of intensity data per pixeland therefore had no analog-like gray scaling, i.e., variable intensity,capability.

The capability of processing a large number of waveforms is a highlydesirable feature in a digital oscilloscope. U.S. Pat. No. 5,530,454 toEtheridge et al. for “Digital Oscilloscope Architecture For SignalMonitoring With Enhanced Duty Cycle”, herein incorporated by reference,describes an oscilloscope that is capable of acquiring up to 400,000waveforms per second when each waveform record is short and thefrequency of trigger events is high enough. This oscilloscope somewhatemulates analog oscilloscope performance with a “live time” percentagethat varies from about 6% to nearly 99%, depending on the timebasesettings, number of channels turned on, and trigger event availability.

The speed of the architecture described in the Etheridge ‘454 patent wasattained by using two raster memories, one in the acquisition system andthe other in the display system. The first of these raster memories,referred to as the “raster acquisition memory”, was almost constantlysupplied with acquired waveforms. These waveforms were rasterized andcomposited into this raster acquisition memory nearly as quickly as theywere acquired by a high speed acquisition rasterizer and similarly rapidimage combiner. After many waveforms had been thus composited, thecontents of the single-bit per pixel raster acquisition memory wastransferred to a multiple-bit-per-pixel display raster memory, where itwas combined with previously displayed data. The contents of thatdisplay raster memory were subject to a digital persistence control,such as that described above, and the intensity of each pixel wasthereby decayed over time.

For many users, especially those having some experience with analogoscilloscopes, variable brightness usefully communicates informationabout the activity of the signal being observed. Many of these usershave had a strong preference for some behaviors that resemble those ofanalog oscilloscopes. For example, as an analog oscilloscope generatesvertical excursions during a horizontal sweep interval to provide areal-time picture of the signal activity at the probe tip, theyinherently tend to vary the brightness of the display as an inversefunction of the slope of the line they produce. This occurs because thecathode electron gun of the CRT generates a constant supply of electronsthat depends on the setting of a “brightness” control, and the length ofthe trajectory covered in a unit of time is minimally determined by thex-axis distance associated with any particular sweep speed, but isincreased by any and all y-axis excursions. And a y-axis excursion canbe a large multiple of the corresponding x-axis distance, so theconstant available electron beam energy can be reduced by a large factoras it is spread over this much longer distance. Thus, analogoscilloscopes inherently vary the brightness of the line they draw as aninverse function of the slope of that line.

Another even more highly desired feature of an analog oscilloscope or adigital oscilloscope with a high waveform throughput, is the ability todetect an intermittent signal anomaly that occurs in an otherwiserepetitive signal. Older digital oscilloscopes, with low “live time”make observing intermittent signal activity improbable, at least in theabsence of special trigger modes designed to detect certain classes ofintermittent signal activity. Analog oscilloscopes will show a fainttrace indicating the presence of this intermittently anomalous signalbehavior. Of course, if the signal becomes too intermittent, the tracewill be so faint in brightness that it may be missed entirely by theoscilloscope operator.

Sometimes, trigger rates are variable in a way the prevents fullutilization of the acquisition and rasterization circuitry. What isdesired is to be able to operate this high speed and high volumeacquisition and rasterization apparatus in a suitable and efficientmanner in the presence of low frequency input trigger rates whilemaintaining a high probability of capturing the data associated withthose slow triggers

BRIEF SUMMARY OF THE INVENTION

In accordance with the invention, a method is provided to enable thishigh speed and high volume acquisition and rasterization apparatus tooperate in a suitable and efficient manner in the presence of low inputtrigger rates. When the expected duration of a waiting interval for theoccurrence of the next trigger is exceeded by a variable factor of,e.g., five, ten, or fifteen, the current acquisition process isterminated and the results are passed on after rasterization has beenallowed to complete normally.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

FIG. 1 is a simplified block diagram of the flow of acquired data as itmoves through the acquisition memory and rasterization section and intothe display section of an oscilloscope according to the presentinvention.

FIG. 2 is a block diagram showing how the rasterization process operateson data supplied by the acquisition memory and produces data for theacquisition raster memory.

FIG. 3 is detailed block diagram showing the inputs, outputs, andsubparts of the counter and latches block shown in FIGS. 2 & 4.

FIG. 4 is block diagram of a particular preferred embodiment of thevariable intensity rasterizer.

FIG. 5 is a simplified block diagram showing where in the displaypreparation process intensity mapping and infrequent events emphasis areapplied.

FIGS. 6A & 6B are block diagrams showing two ways that non-linearitycorrection can be performed.

FIG. 7 is a schematic diagram of the intensity mapper circuit foraccomplishing bit-per-pixel reduction in a manner in which the scalingand gain is controllable.

DETAILED DESCRIPTION OF THE INVENTION

Referring first to FIG. 1, we see a simplified block diagram of the flowof acquired data as it moves through the acquisition memory andrasterization section 200 and into the display section 300 of anoscilloscope according to the present invention. Vertical scaling andvertical offset functions are applied prior to digitizing, and aretherefore not directly visible in this figure, but rather are part ofthe acquisition process-10. The trigger circuitry 15, under thedirection of the process controller 180, monitors the input signal andother criteria (not shown) and communicates the occurrence of thesatisfaction of the trigger conditions to the acquisition circuitry 10and the process controller 180.

The process controller 180 contains timers and state machines that allowit to control all of the acquisition and rasterization process,including some of the optional processes that will be discussed belowand are not shown in FIG. 1. As can be seen, it sends command andinformation to, and receives various signals from, almost every part ofthe acquisition memory and rasterization section 200 as well the trigger15 and acquisition circuitry 10. It notifies the raster combiner 80 whendata is ready. The process controller 180 monitors the trigger circuitry15 and starts the acquisition circuitry 10 and the rasterizer 30 atappropriate times, i.e., when there are enough acquisitions to rasterizeor else so much time has elapsed from the first trigger or the lastdisplay update that new data is due at the display 110.

When data is due at the display 110, the process controller 180 startsthe intensity mapping 50 and subsequently the DMA circuitry 70. As willbe discussed in more detail below, the process controller 180 alsodetermines when acquisition aborts are necessary, causes a triggerposition calculator to run, and computes breakpoints.

As the output of the acquisition process 10, voltage-versus-timedata-address pairs are stored in an acquisition memory 20. Acquisitionmemory 20 can hold two waveform records that are up to 512K sampleslong, or may be partitioned to hold up to 256 shorter waveform recordseach containing 768 samples. Each sample location contains 8 bits ofinformation defining one of 256 possible voltage amplitude levels. 200of these voltage amplitude levels correspond to one of the 200 pixellocations in each column of the 21-bit-per-pixel 200×500 acquisitionraster memory 40.

Each acquisition memory and rasterization section 200 can acquire100,000 waveform records per second, with each record containing 500data points, and with each data point acquired at 1 ns intervals, for atotal of 500 ns per acquisition. Multiple acquisition memory andrasterization sections 20, specifically two or four, can be interleavedtogether to double or quadruple the overall throughput available on oneoscilloscope channel. The raster combiner 80 of the display section 300can multiplex the outputs of these multiple acquisition memory andrasterization sections 20 back together in the display section 300.

Conversely, more than one channel can be multiplexed into oneacquisition memory and rasterization section 200. And an alternativeversion of the raster combiner 80 can demultiplex these multiplechannels out of fewer acquisition memory and rasterization sections 200and, through the use of extra “tag” bits in each pixel location of thedisplay raster memories 90 and 100, cause them to be displayed asseparate channels on the raster display 110. The tag bits allow thechannels to be prioritized, or “layered”, so that only the intensity ofthe one on top is displayed when two or more channels overlap.Alternatively, the intensities from two or more channels can be addedtogether when they overlap, if so desired.

The raster combiner 80 can also be made to translate intensityvariations into color variations, with appropriate supportive changesbeing made in the display raster memories, 90 and 100, and the rasterdisplay 110, if needed.

Referring next to FIG. 2, please note that, for simplicity's sake, clockand timing signals, and control signals from the process controller 180are not shown in this figure. The rasterization process 30 accessesappropriate samples from the acquisition memory 20 according to theaddresses generated by the address controller 31. The address controller31 receives time-per-division settings and is preprogrammed with part ofthe information shown in Table 1. It is thereby able to convert thosetime-per-division settings into addresses that reflect the desiredcombination of compression width and acquisition depth that will beincluded in the determination of the intensity for each pixel. The restof the contents of Table 1 are known to the process controller 180.

The address controller 31 fetches acquisition data as directed by theprocess controller 180 (shown in FIG. 1), while providing sequentialsample offsets and buffer selection information to the data buffer 32.The address controller 31 also identifies the end of each column so thatthe counter block 36 can be reset and latched appropriately. Thesecontrols, in effect, pass alongside the data pipelines so that thecontrol tracks the flow of the data. When the raster updating takeslonger than the compression process, the compression process issuspended at the end of the then current compression operation.

TABLE 1 TIME/DIVISION PROGRAMMING Index # Time-Per-Division Count SizeUsed E.T. or Decimation Expand/Compress  0 200 ps 256 768 2 Equiv. Time= ON Expand = 250  1 250 ps 256 768 2 or 3 Equiv. Time = ON Expand = 200 2 500 ps 256 768 5 Equiv. Time = ON Expand = 100  3 1 ns 256 768 10Equiv. Time = ON Expand = 50  4 2 ns 256 768 20 Equiv. Time = ON Expand= 25  5 2.5 ns 256 768 25 Equiv. Time = ON Expand = 20  6 5 ns 256 76850 Equiv. Time = ON Expand = 10  7 12.5 ns 256 768 125 Equiv. Time = ONExpand = 4  8 25 ns 256 768 250 Equiv. Time = ON Expand = 2  9 50 ns 128784 500 No decimation Compress = 1:1 10 100 ns 77 1312 1000 Nodecimation Compress = 2:1 11 200 ns 31 1312 1000 Decimate by 2 Compress= 2:1 12 500 ns 16 3264 2500 Decimate by 2 Compress = 5:1 13 1 μs 8 32642500 Decimate by 4 Compress = 5:1 14 2 μs 4 3264 2500 Decimate by 8Compress = 5:1 15 5 μs 2 6336 5000 Decimate by 10 Compress = 10:1 16 10μs 1 6336 10K Decimate by 20 Compress = 10:1 17 20 μs 1 12688 20KDecimate by 20 Compress = 20:1 18 50 μs 1 25392 25K Decimate by 20Compress = 50:1 19 100 μs 1 50800 50K Decimate by 20 Compress = 100:1 20200 μs 1 101552 100K Decimate by 20 Compress = 200:1 21 500 μs 1 256000250K Decimate by 20 Compress = 500:1 22 1 ms 1 512000 500K Decimate by20 Compress = 1000:1 23 2 ms 1 512000 500K Decimate by 40 Compress =1000:1 24 5 ms 1 512000 500K Decimate by 100 Compress = 1000:1 25 10 ms1 512000 500K Decimate by 200 Compress = 1000:1 26 20 ms 1 512000 500KDecimate by 400 Compress = 1000:1 27 50 ms 1 512000 500K Decimate by1000 Compress = 1000:1 28 100 ms 1 512000 500K Decimate by 2000 Compress= 1000:1 29 200 ms 1 512000 500K Decimate by 4000 Compress = 1000:1 30500 ms 1 512000 500K Decimate by 10000 Compress = 1000:1

Table 1 contains the values used for different time-per-divisionsettings. The leftmost column shown in Table 1 contains an index numberthat allows reference to particular rows and the sets of data that theycontain. The next column, “Time-per-Division”, contains an entry forevery horizontal setting of the first oscilloscope embodying the presentinvention. The “Count” column indicates how many acquisition records areused as input to each rasterization process at the corresponding“Time-per-Divison” setting. In equivalent time operation (furtherdescribed below), only one of the 256 records is processed during eachrasterization cycle. However, the number of buffers in the memory, 256,is twice the number, 128, in the only comparable non-E. T. setting,index #9, 50 ns per division. The “Size” column indicates how many datapoints are stored in each rasterization record at this setting, whilethe “Used” column indicates how many data points are actually used inperforming the rasterization.

The column, “E. T. or Decimation”, contains information referring to theamount of “equivalent time” filling-in or decimation (effectively“throwing out”) that will be performed by the acquisition hardware.Equivalent time sampling is a technique for effectively acquiring datathrough multiple acquisitions that corresponds to a sweep speed orsample rate that is actually faster than the acquisition hardware canperform. In the acquisition record produced by equivalent time sampling,anomalies may appear that result from the hypothetical nature of theacquisition record produced by the non-sequential acquisition of“successive” data points and the fact that the waveforms sampled toproduce this record may not be perfectly repetitive.

The next-to-the-last column, “Expand/Compress”, contains informationthat refers to the amount of expansion or compression that will beperformed during the rasterization process. It is the combination ofthese factors, expansion and compression, and those from the precedingcolumn, equivalent time filling in or decimation, that determine whichsamples contribute to the record in the acquisition memory 20 andultimately which samples affect the rasterization result sent to theacquisition raster memory 40. For a single acquisition memory andrasterization section operating in its single channel mode, 6% of realtime is spent acquiring, as compared with 94% spent processing theacquired acquisitions.

Index row #9, the 50 ns time-per-division row, contains the fastestreal-time settings. All of the rows above row #9, those withTime-Per-Division settings faster than 50 ns per division, require theuse of equivalent time sampling, i.e., the collection of multiplerecords and the use of only some data points from each one to build acomposite data record. For instance, in row #0 the 200 ps per divisionis achieved by using only two data points each from 250 acquisitions.The oscilloscope is acquiring samples once every nanosecond, so only twodata points per record are within the time frame of one record. At 200ps per division on the horizontal timebase, and 10 divisions perdisplay, each display is only 2 nanoseconds from side to side. Bycombining two samples each from 250 records, 500 records per display,can be composited into one equivalent time image.

In the rows below index #9, decimation is used to limit the recordlength and compression is used to shorten the distance taken up on thescreen by the points contained in the record. For example, at 500 ms perdivision, the 10 division wide screen represents 5 seconds. At the onesample per nanosecond sample rate, 5 billion samples are availableduring that 5 seconds. Decimation by 10,000, allows 500,000 data pointsto be collected. Compressing those 500,000 data points with acompression factor of 1000, produces 500 points per display and 50points per division.

Referring again to FIG. 2, the data from the acquisition memory 20 thatis specified by the address controller 31 is transferred to the databuffer 32. While the data buffer 32 can hold a total of 64×16 bytes ofdata, it actually receives only 8 bytes at a time every 16 ns. While thedata buffer 32 is supplying one set of 8 bytes of acquisition data tothe sample to vector converter 33, the other 8 bytes are being filled bythe next transfer from the acquisition memory 20. 8 bytes per 16 nsequates to an average of 2 ns of access time per single sample point ofacquired data.

The 8-bit data from acquisition memory 20 defines 256 vertical voltagelevels, while the acquisition raster memory has vertical columnscontaining 200 pixels. Thus, there are 56 extra voltage levels definedby those 8-bit outputs which do not correspond to any of the 200vertical pixel locations. Of those, 28 represent voltages below thebottom of the 200 vertical pixel locations corresponding to the displayscreen and 28 represent voltages that are above the 200 vertical pixellocations and the top of the screen. The screen typically shows agraticule with nine horizontal lines defining 8 vertical divisions ofthe display, each of which will receive data from 25 of the 200 pixelsin each column of the acquisition raster memory 40. As will be furtherexplained below, a one-to-two vertical expansion will occur before thisdata reaches the actual screen, so ultimately each vertical (andhorizontal)division of the display will correspond to 50 pixels.

While the 28 top and 28 bottom values do not directly affect the 200pixels in each column, as will be further explained below, they do exertan indirect effect by being used in the calculation that determines theslope of those vectors that will affect the 200 pixels in the presentcolumn. Affecting the slope calculation of a vector exerts an effect onthe incremental intensity values that the pixels in that vector willreceive.

The 28 top and bottom values also have one other effect. If a“clip-to-screen” function is active, and all of the current vector is inthe top 28 or bottom 28 vertical locations, i.e, all of the points inthe current column are off the screen in one direction or the other, aspecial tag bit will be set in the 1st or 200th location in theacquisition raster memory 40 to indicate that condition.

Continuing with FIG. 2, the sample-to-vector converter 33 examines threesample data points, N, N-1, and N-2, in connection with producing eachvector Top and Bottom signal. If all signal data were monotonic, twodata points would be sufficient. However, the possibility ofencountering inflection points in the data, necessitates the use ofthree points. The sample-to-vector converter 33 supplies a Hi and Losignal to absolute value subtractor 34. The Hi and Lo signals can have amaximum difference of 255 vertical increments, since they are derivedfrom the full span of values potentially defined by each 8-bit datasample. Absolute value subtractor 34 produces a vertical (or voltage)change value, dV, that it forwards to look-up table 35. Look-up table 35uses the vertical change value, dV, to find an appropriate weight, W,for the present vector. In the presently described implementation, W canhave values from 0 to 31. Generally, W has a value that bears an inverserelationship to dV/dT. For a particular time-per-division setting, dTwill have a constant relationship to the distance represented by eachcolumn of pixels to be rasterized. For more information on how W may becalculated, refer to U.S. Pat. No. 5,550,963 to Siegel et al. for“Graded Display of Digitally Compressed Waveforms”, hereby incorporatedby reference.

Sample-to-vector converter 33 also supplies a pair of Top and Bot(tom)signals to the 200 counter block 36. These signals define, respectively,the values from 1 to 200 that define the length of the unsigned vectorto be drawn as a result of the current sample data point. Each counterin the defined region is incremented by the value of W, the vectorweight output of look-up table 35. It should be pointed out that the 8bits of voltage amplitude data associated with one data point undergoesa potentially huge expansion during the process of rasterization. Sincethe location that it specifies can be up to a full vertical screendistance away from the location specified by the preceding data point,as many as 200 pixels can require updating as a result of thevectorization of one acquired data point. Since there are 19 bitcounters and 4 tag bits, and these are later compressed into 21 bits ofinformation at each of those 200 pixel locations, the totalrasterization process for a single data point of 8 bits can affect thestatus of as many as 4,200 bits in the 21-bit acquisition raster memory40. It thus becomes important to keep this data expansion fromunnecessarily impeding the waveform throughput of the acquisition memoryand rasterization section 200, and thereby that of the whole digitaloscilloscope (given the availability of a limited number of suchsections).

Referring now to FIG. 3, the 200 counter block 36 and latch 37 are shownin detail, along with their inputs and outputs. The left side portion ofFIG. 3 is a bank of 200 4-bit channel tag registers 36B. The centralportion of FIG. 3 is an array of 200 19-bit counters 36A. The 19-bitlength of the counters in the counter block 36 is chosen in part forcompatibility with the use of this same circuitry in anotherapplication. Because the maximum value of W is 31, and 2¹⁹/31=16912, themaximum compression factor resulting from this depth is 16,000.

Top and Bot(tom) input signals are each 8-bits long and identify one ofthe 200 19-bit counters. Weight, W, is a 5-bit signal that specifies thevalue from 0 to 31 by which each 19-bit counter in the vector defined byTop and Bot(tom) are to be incremented. The right side portion of FIG. 3is a bank of 200 23-bit latches 37. These 200 23-bit latches 37 hold foroutput the contents of the both the 200 4-bit channel tag registers 36Band the array of 200 19-bit counters 36A, while those registers andcounters are cleared (using the clear signal) and used to accumulate thenext column of data. This array of counters is relatively expensive toimplement and is typically able to draw vectors at a slower rate thansample data can be accessed from an acquisition memory. By including thelatches 37 with these counters, it is possible to update a currentcolumn while drawing the next column into the counters, thereby avoidingany underutilization of this resource.

The 19 bits of intensity data in the bank of 200 23-bit latches 37 isadded to the existing contents of the appropriate column of theacquisition raster memory 40 by adders 38, in a series ofread-modify-write operations, as addresses are provided by rasteraddress generator 45. The 4 bits of channel tags are OR-ed withcorresponding bits from the acquisition raster memory 40, during thesame read-modify-write operations. The channel identification bits, ofwhich there are only enough to describe the number of channelsmultiplexed into one acquisition memory and rasterization section 200,are packed into the 21-bit pixels by sacrificing MSB's of intensitydescription capability.

When priority coding is used to determine the intensity at regions ofchannel overlap, there is lower throughput per channel, but a smallernecessary dynamic range. The worst case occurs when regions of overlapare added together for four channels (instead of the priority of thechannels determining which intensity is displayed). There are then only17 bits available for intensity information and they are saturatedsooner because the overlapping intensities add. Since the maximumintensity update weight, W, is 31, the smallest number of updates thatcan produce saturation is 2¹⁷/31=4338. When acquiring 100,000 waveformsper second, needing an update every 4338 acquisitions implies a minimumrequired update rate of 23 times per second. Since the normal updaterate will be between 30 and 60 times per second, this requirement willusually be easily satisfied. The channel identification is packed intothe 21-bit pixels throwing away MSB's of intensity capability. Only thebits necessary to deal with the number of channels rasterized in oneacquisition memory and rasterizer section 200 are sacrificed.

Acquisition raster memory 40 requires 268K bytes of memory to support a200×500 array of pixels. 8 bytes of this memory are accessiblesimultaneously. Therefore, each access contains 21 bits of data for eachof 3 pixels (with 1 bit wasted). A column of 200 pixels requires 67accesses of 3 pixels each. Thus,8(bytes/access)×67(accesses/column)×500(columns/display)=268K(bytes/display).

Referring now to FIG. 4, in an actually constructed embodiment, it wasfound to be efficient to combine acquisition memory 20 and acquisitionraster memory 40 in one large combined acquisition memory 20,40. Also,while until this point in the discussion, the description has beenlimited to conventional voltage-versus-time, or “YT”, displays, it canbe seen in FIG. 4 that other modes, such as “XY” and “XYZ”, can also beintegrated into one overall design. Address source selector 46 has asone of its sources ET/XY/XYZ addressing source 41, as well as rasteraddress generator 45 and the two parts of address controller 31 (of FIG.2) shown here as 256×16 RAM 31A and 2×8 pointer extension 31B. Thepointer extension is supplied for rasterizing a single, >65536 buffer.Since the rasterization process operates on either a lot of smallbuffers, a single large buffer, or something in between, a tradeoff ismade on wasting space with RAM vs. length and count that can berasterized at one time.

Continuing the process of relating what is shown in FIG. 4 with what hasbeen discussed above in FIG. 2, sample processor 33′ corresponds tosample-to-vector converter 33. FIG. 4 does not show the distinctionbetween the Top and Bot(tom) signals and the Hi and Lo signals that wasmade in FIG. 2, but does show that the Bottom signal is one of thepotential inputs to addressing source ET/XY/XYZ 41 through multiplexer44, which has as its other input “selected A/D data”. “Selected A/DData” is used in XY and XYZ modes, the data processor, acquisition databuffer, delta to intensity map, and acquisition queue pointers are notused in XY and XYZ modes. The delta-to-intensity map 34,35 and thecounter block 36 are not used during equivalent time (ET) operations.

Adders 38 shown in FIG. 2 are shown in more detail in FIG. 4 as threeinstances of 21-bit saturating adders. Saturating adders do not overflowor produce MSB carries, but instead remain at a maximum value of all 1'swhen they become full. Read and write access to the (combined)acquisition memory 20,40 each take 16 ns and are 64 bits wide.Therefore, a Read-Modify-Write cycle can be performed on three 21-bitpixels every 32 ns. That is an average time of 10.67 ns per pixelupdate. Therefore, the longest that it can take to update all 200 pixelsin a column is 67×32 ns or 2.144 μs. Thus, if the input data for eachcolumn includes 134×8/7=154 data points from the acquisition memory 20,the acquisition raster memory 40 update time will not limit therasterization speed and the rasterization process speed itself becomesthe limiting factor. Since the (combined) acquisition memory 20,40 canaccessed to provide 8 bytes in parallel every 16 ns, it is possible toread acquisition waveform data 8 times faster than it can be processed.Because the data buffer 32 can hold up to 128 8-byte acquisitionsegments for parallel processing, 87.5% of the (combined) acquisitionmemory bandwidth is available for updating the acquisition raster memorywith the output of the rasterization process.

If, as in this design, the number of pixels per column is 200(M=200pixels/column), and the average time required to update each pixelis 10.67 ns (P=10.67 ns/pixel-update), the throughput rate of therasterizer is 2.134 μs/column (M*P=2.134 μs/column). Then, if the timeto draw the vector based on a single new data point is 16 ns (D=16ns/sample-pair), and the average time required to access a single samplepair of acquisition data is 2 ns (A=2 ns), the available time forupdating the acquisition raster memory is D-A, or 14 ns. Thus, themaximum combination N*C, of acquisition pixels per column, N, andcompression pixels within each column, C, that should be used to keepthe rasterizer fully occupied (assuming the availability of sufficienttriggers) is: N*C=(M*P)/(D-A), or N*C=(2134 ns/column)/14 ns, orN*C=152.4 pixels. And, if the appropriate compression factor, C, isknown, N can be calculated according to: N>152.4/C. This efficiencyfunction is specific to the design shown in FIG. 4, where the outputbandwidth and input bandwidth share a single overall memory bandwidth,i.e., A subtracts from D.

A more general efficiency function arises in connection with FIG. 2,where the input and output bandwidths are independent of each other. Inthat case, the ideal number of acquisitions N, when C, P, and M aredetermined, is: N=(M*P)/(C*D). By designing and programming anoscilloscope rasterizer in this way, it is possible to supply images tothe screen at a constant samples/sec rate at all time/divs when thereare a sufficient number of triggers. In order to maintain a uniformacquisition rate, it is important to have twice as many acquisitionbuffers as are used in a single pass of rasterization. In this way,acquisitions may continue into the buffers which are not currently beingrasterized. These buffers may either be arranged as two banks that arealternately acquired and rasterized, or as a circular queue of bufferswith rasterization starting on a display update deadline or when thedesired number of records have been acquired.

Referring again to FIG. 1, the contents of 200×500 21-bit-per-pixelacquisition raster memory 40 is then “mapped” 50 into a second, shorterdepth, 4-bit-per-pixel acquisition raster memory 60 having the sameplanar (200×500) dimensions. This shorter 4-bit-per-pixel acquisitionraster memory 60 still contains 200×500 pixel locations, but each pixelonly has 4-bits-per-pixel of intensity information associated with it,and consequently can only display 16 levels of intensity. This “mapping”50 is accomplished by the use of 15 breakpoints that (monotonically)define the boundaries of the 16 intensity levels. In one embodiment, themapper 50 is configured to process three 21-bit pixel value inputs inparallel every 16 ns. This approach requires that there be severalpipeline delays between the 16 ns read of three 21-bit pixel values andthe corresponding 16 ns writes of five groups of three 4-bit pixel valueoutputs into the 8-byte wide memory. Because the mapping operationdepends on using data stored with in the acquisition raster memory 40,exporting that data is another constraint on the overall utilization ofthat limited bandwidth. Exporting the data for the mapping uses about600 μs for each display update or about 0.6 ms/30 ms. This results in a2% drag on the efficiency of the rasterization process.

The contents of the short 4-bits-per-pixel acquisition raster memory 60are transferred to a raster combiner 80 in the display section 300 by adirect memory access (DMA) process 70. The display section 300 containstwo display raster memories 90 and 100. These raster memories can alsobe 4-bits-per-pixel, or can include additional bits for channelidentification and coloring. Four tag bits can identify up to fourindividual channels or 16 combinations of channels.

While one 90 of two display raster memories 90 and 100 are supplying theraster display 110 with data for the current display, its contents arealso being combined with the contents of the short 4-bits-per-pixelacquisition raster memory 60 by the raster combiner 80. The output ofthat process is stored in the other display raster memory 100. Duringthe vertical retrace time of the display 110, the two display rastermemories 90 and 100 are functionally swapped, and the raster combiner 80output is then stored in the opposite display raster memory 90 while itsinput and that of the raster display 110 are provided display rastermemory 100. During the raster combining process, the 200 verticallocations of the acquisition raster memories are expanded to 400 suchlocations in the two display raster memories 90 and 100. In a simpleembodiment, this is accomplished by duplication of each line of data,although more sophisticated means, such as averaging a line above and aline below, could be employed if so desired.

While in one implementation each acquisition memory and rasterizationsection 200 only handles data associated with one channel of theoscilloscope, in an alternative embodiment it is possible to put datafrom multiple channels into a single acquisition memory andrasterization section 200. When this is done, samples from differentchannels are interleaved in each “acquisition” memory record, so thateach sample from a particular channel is a constant and predeterminednumber of locations apart in the data record. Harking back to theterminology of analog oscilloscopes, in which it was possible to have a“chopped” view of two or more signals displayed by the multiplexed useof a single horizontal timebase, this way of using the acquisitionmemory and rasterization sections 200 is referred to as “chop” mode.When data from different channels is interleaved in the chop mode, thedata associated with individual channels is “tagged” by therasterization process 40 in extra bits associated with each pixel, sothat it may later be determined which channels have affected thecontents of each pixel. In the 200 counter block 36 there are four extrabits for channel tagging besides the 19-bits that are part of thecounters.

To maintain maximum throughput in the system described above,acquisitions are processed in batches. To keep the flow of acquisitiondata constantly available to the rasterization process at the necessaryspeed, the next acquisition must be started after the average singleacquisition rasterization time has elapsed since the end of the previousacquisition. However, acquisitions start by sampling data and moving itinto the acquisition memory, without knowing when, or even if, the nexttrigger will ever occur. Yet, the goal of the acquisition memory andrasterization section 200 is to keep data flowing toward the displaysection 300 without any perceptible interruption. Therefore, therasterization process 30 cannot engage in indefinite waiting for atrigger to be received, if its ability to continue outputting data isrendered impossible by that course of action.

To deal with this problem, the process controller 180 keeps track of thetime that passes after the next acquisition should have become availablebut has not done so. When such a “late trigger interval” has elapsed,the process controller 180 queries the trigger circuitry 15 to see if atrigger has been received yet. If a trigger has been received, there isno point in abandoning an acquisition that will soon be completed in anyevent. However, if no trigger has been received, and the processcontroller 180 does not receive a “trigger received” signal from thetrigger circuitry 15 in response to its query, it “aborts” the currentacquisition process and forwards the partial results to the intensitymapping circuitry 50 (via the non-linearities correction circuitry 130if present). Typically, the late trigger interval is on the order of 10times the length of time between successive acquisitions. This permitsslow triggers with a uniform distribution to be detected about 90% ofthe time. This solution creates another problem in the presence ofperiodic slow triggers arriving at an interval that is about equal tothe normal acquisition delay plus the late trigger interval. To avoidmissing almost all these triggers, the late trigger interval can be madeto vary randomly over some period of time, e.g., from 50% to 150% of the10 times the normal acquisition rate number.

The design described above, with its deep statistical database of the21-bit-per- pixel acquisition raster memory 40 and the high through-putvariable intensity rasterization process 30 that supplies it with data,makes visible artifacts that are not seen in prior art rasterizers.Specifically, A/D differential non-linearity (A./D DNL) from theacquisition process 10 can result in horizontal banding becoming visiblein the display 110. Also, when timebase selections require the use ofequivalent time settings, time interpolator non-linearities (TINL) canshow up in the display 110 as vertical banding.

Referring first to FIG. 5, non-linearity correction circuitry 130 can beused to allow compensation for these factors as the pixel intensity datamoves from the acquisition raster memory 40 to the intensity mappingprocess 50. Referring next to FIGS. 6A and 6B, individual pixelintensities can be successively multiplied by correction coefficientsfor the A/ID differential non-linearities with multiplier 131 and thetime interpolation non-linearities with multiplier 132, as shown in FIG.6A, or the A,D DNL and TINL correction coefficients can be multiplied byeach other first 133 and then the combined correction coefficients canbe applied to the pixel intensities by multiplier 134 as shown in FIG.6B.

Returning to FIG. 5, the correction coefficients are supplied to thenon-linearities correction circuitry 130 by correction look-up table150. As the correction coefficients, both A/D DNL and TINL, may bothvary on a pixel-by-pixel basis, the correction look-up table 150 isaccessed by raster addresses from the raster address generator 45 shownin FIG. 2 (with suitable delay, if needed). If linearity correction isemployed, the input to the intensity mapping circuitry 50 will be fromthe output of multiplier 132 of FIG. 6A or from that of multiplier 134of FIG. 6B, one or the other of which is shown as the non-linearitycorrection block in FIG. 5.

The correction coefficients needed for the correction look-up table 150can be determined at the factory or by built-in instrument calibrationsoftware. If the DNL and TINL are stable with instrument age andtemperature, then a set of factory determined values would besufficient. Otherwise, instrument signal path compensation softwarecould do the calibration at user request when enhanced accuracy isdesired.

Determining the A/D DNL compensation coefficients could be accomplishedby supplying an ideal sine wave or some other known ideal waveform tothe A/D, digitize it, and accumulate the statistics for hit rate foreach digitizing level. The acquired statistics for the proportion oftime that the waveform was at each voltage level can be compared toknown statisitics for the ideal waveform to determine suitablecorrection coefficients to make the statistics match.

To determine the correction coefficients for the time interpolatornon-linearity, a fast trigger, asynchronous to the scope, should besupplied to the trigger circuitry, and a histogram profile of thedistribution of the hits compared with the uniform average level thatwould be expected of an ideal perfectly linear time interpolator. Again,suitable compensation factors would bring the actual data intoconformity with the ideal.

Performing a linear intensity accumulation of the acquired data,according to the rasterization method described above, means that thegeneral level of accumulated intensities is directly proportional to thenumber of acquisitions accumulated. This raster is periodically clearedor decayed each time the screen is updated. If the time between updatesis not constant, as it frequently may not be due to other processoractivity, other display activity, or data dependent raster copyingtimes, a fixed scaling of the accumulated acquisition raster memorycontents can lead to observable, non-signal related intensityvariations. Also, if an exponential decay is applied to the accumulatingacquisition raster memory contents between display updates, there willbe an initial transient build up of overall intensities in the map untila status quo is reached. These effects are all unwanted artifacts thatit would be desirable to eliminate.

While a linear accumulation of acquisitions into intensity variations isthe easiest to produce at the rasterizer end of the process, an optimalpresentation from the user's perspective might be more non-linear. Forexample, the user might sometimes be strongly focused on the mostinfrequent events and therefore might want to have all non-zero pixelsmapped to the brightest intensity that the display can produce.Alternatively, there might be a certain region of pixel hit rates, andcorresponding accumulated intensity, that the user would like to map tothe full range of dynamic intensity available in order to more clearlydistinguish subtle differences in the frequency at which they are beingaffected by acquired data.

Both of the above sets of goals can be addressed with a usercontrollable intensity mapping function that allows control of the gainand offset of the transfer curve. From the user's point of view suchcontrols could appear, respectively, as contrast and brightness. Themeans by which the transfer function can be modified, when it isimplemented as shown and described in FIG. 7 below, is by controllingthe magnitude of a maximum pixel intensity reference value andcontrolling the size of the fifteen break point fractions that make upthe mapping function. Each of the fifteen break points is determined bymultiplying the corresponding fraction times the maximum pixel intensityreference value. The fractions define a normalized transfer functionthat the user can modify, while the maximum pixel intensity valueprovides scaling for the expected data accumulations in the acquisitionraster memory.

Referring now to FIGS. 5 & 7, the intensity mapping process 50 iscontrolled by break point values. FIG. 7 shows how these break pointsare used in the intensity mapper circuit to accomplish bit-per-pixelreduction in a desired manner. The assignment of 15 break point valuesdetermines the boundaries of the 16 intensity levels that will beproduced by the 4-bit-per-pixel mapping.

The circuitry shown in FIG. 7 operates as a successive approximationdigitizer as it reduces the bit count of the pixels. First, however, the21-bit pixel intensity word must be trimmed to eliminate any tag bits.As tag bits, if present, will be the most significant bits (MSBs), andthere will at most be four of them, the four MSBs are ANDed with a 4-bittag mask by AND gates 51. The TagBits/Value table 59 shows how thenumber of tag bits present, shown in the left column, are masked out bybits representing the hexadecimal value shown at the right, therebyproducing 17 to 21 bits that are all representative of the input pixelintensity value.

The input pixel intensity value is compared 52 with Break Point 8, whichhas a value corresponding to the MSB of the number of bits in the maskedinput pixel intensity value, i.e., one half of the maximum input pixelintensity value. If the input pixel intensity value is larger than thevalue of Break Point 8, Bit 3 (the MSB) of the mapped-to 4-bit pixelintensity value will be one. Conversely, if the input pixel intensityvalue is smaller than the value of Break Point 8, Bit 3 (the MSB) of themapped-to 4-bit pixel intensity value will be zero. In similar fashion,the input pixel intensity value is compared 54 with either Break Point 4or Break Point 12, representing ¼ and ¾, respectively, of the inputpixel intensity value range to determine Bit 2 of the mapped-to 4-bitpixel intensity value. Whether the compared-with value is ¼ or ¾ dependson which output of multiplexer 53 is selected by the outcome of theprevious comparison 52.

Similarly, the results of the first two comparisons 52,54 select whichoutput of multiplexer 55 the input pixel intensity value is to becompared 56 with, Break Point 2, 6, 10, or 14, representing,respectively, ⅛, ⅜, ⅝, or ⅞ of the maximum input intensity value. Theresult of this comparision 56 is Bit I of the mapped-to 4-bit intensityvalue. This process is repeated one additional time, with the output ofmultiplexer 57 being one of Break Points 1, 3, 5, 7, 9, 11, 13, or 15,depending on the outcome of the first three comparisons 52,54,56, andwith that output being compared 58 with the input pixel intensity valueto determine the state of Bit 0. The output of this process, which canalso be characterized as a binary search to locate mapped-to bins, isthe 4-bit pixel intensity value.

While the input pixel intensity values and the break points are normallyall 21 bit integers, if there are tag bits, the MSBs of the input pixelintensity values are forced to zero and the break points are made to berestricted in range. This scales all of the break points down to being afraction of the reduced-in-value maximum pixel intensity referencevalue.

Other implementations than the one shown in detail above could used toperform the actual mapping. A hardware or software implemented lookuptable, or software algorithm implementing a binary search or some othersorting function could also be used to convert input pixel intensityvalues into output pixel intensity values based on the break points andfractions that they represent. The most important aspect and real valueof the invention is in being able to vary the magnitude of the maximumpixel intensity reference value in various ways and also in being ableto variously define the break points and thereby shape the transferfunction between the two pixel intensity representations. The number ofinput bits and output bits can be varied, so long as the number ofoutput bits is smaller than the number of input bits. The number N ofpossible output intensity values could be other than a power of two. Thenumber of fractions and break points used to implement the transfercould be less than N-1, thereby altering and partially collapsing thesymmetry of the mapping function.

There are also a number of interesting possible ways to establish themaximum pixel reference value, in addition to simply determining themaximum pixel intensity value that can be represented by the greaternumber of input bits. The operator could be permitted to set the maximumpixel intensity reference value through the use of a “brightness”control. The alternative actually implemented allows the operator toselect between an “Autobrightness ON” mode and an “Autobrightness OFF”mode. In the Autobrightness ON mode, the maximum pixel reference valuecan be made a function of the actual intensity value of the largestinput pixel intensity value, as measured by hardware or softwaresomewhere in the data path. Conversely, in the Autobrightness OFF mode,the maximum pixel reference value can be made a function of an amount oftime that intensity data has been accumulating into acquisition rastermemory 40. While the foregoing use of the Autobrightness mode conceptpresently appears to be the preferred embodiment, the ON mode coulddepend on some factor other than the actual intensity value of thelargest input pixel intensity value, or the OFF mode could depend onsome other factor than time acquired.

The maximum pixel intensity reference value can also be made a functionof the average pixel intensity value of the pixels stored in theacquisition raster memory 40. Alternatively, the maximum pixel intensityreference value can also be made a function of the average non-zeropixel intensity value of the pixels stored in the acquisition rastermemory 40.

In the case of Autobright OFF as it is originally discussed above, themaximum pixel value used is a theoretical maximum pixel intensity valuewhich is directly proportional to the amount of time that acquisitionshave been accumulated into the image. This Autobrite OFF option causesthe overall process to duplicate an artifact of analog oscilloscopes,namely that the display intensity depends on the user's trigger rate. Insome circumstances, this conveys additional information to the user,specifically a qualitative indication of signal rate. In othercircumstances, this behavior may be objectionable, thus the alternativeof Autobright ON and automatically scaling all events to actual fullintensity. The use of a theoretical maximum pixel intensity based onlyon the time used for acquisition can lead to a mismatch of the imagepresented as compared with an image derived from the use of scalingbased on the actual maximum pixel intensity received. Normally, whenplenty of triggers are available, the system would be expected toproduce the same image without regard to which of these references isused. In practice, they may be very different and it is desirable toallow the user to control which is used.

Depending on the maximum slew rate present, a wide dynamic range ofmaximum pixel intensity will occur. For example, the maximum pixelintensity value for a signal that remains at a constant voltage levelwill be 200 times as large as that of a signal that spans all 200voltage levels of the display in the time associated with one pixelcolumn. This variation of overall intensity level will become muchlarger if the dv/dt weighting factor W is being applied. This weightingfactor W produces another factor of 31:1 in intensity difference betweensignals with maximum and no vertical change, making the overallintensity difference factor 6200:1. One method to compensate for thissignal related (as opposed to trigger related) intensity variation is tomultiply the time based theoretical maximum pixel intensity by anefficiency factor. This efficiency factor is based on the ratio of theactual maximum pixel intensity to the expected theoretical maximum pixelintensity, which is itself equal to the compression factor times theacquisition count. In total then, the system computes thetheoreticalMaxPixlntensity=acquistionCount * scaleFactor *(actualMaxPixIntensity / (acquisitionTime * compressionFactor)).

Yet another alternative for the computation of the maximum pixelintensity reference value is to base it on the average non-zero pixelintensity or the average pixel intensity including zero pixels. In oneimplementation the hardware includes means for counting all of thenon-zero pixels and the sum of all pixel intensities, allowing either ofthese approaches to be used.

Referring again to FIG. 5, the output of intensity mapping 50 is nextoperated upon by intensity converter 120 if, but only if, the infrequentevents emphasis enable signal is active. This signal is activated byoperator control when the operator want to emphasize infrequent eventsand de-emphasize more frequent ones. When enabled, intensity converter120 modifies pixel brightness to emphasize infrequent events. In itssimplest implementation this can mean complementing all non-zero pixelintensities. This makes the least bright pixels into the brightest, andthe most bright pixels into the dimmest. Pixels of intermediatebrightness are minimally affected or not affected at all. Other, morecomplex algorithms could be employed using a look-up table or tables.The creation or selection of his table can be under the user's control,and could include mappings that allow the user to highlight by choosingamong frequencies of occurrence, either with intensity or color.

While a preferred embodiment of the present invention has been shown anddescribed, it will be apparent to those skilled in the art that manychanges and modifications may be made without departing from theinvention in its broader aspects. The claims that follow are thereforeintended to cover all such changes and modifications as are permitted bythe patents laws of the respective countries in which this patent isgranted.

What is claimed is:
 1. A method for maintaining rasterizer functionalityin the absence of expected trigger signals, the method comprising thesteps of: determining an expected trigger interval; starting a newacquisition process in advance of when the expected trigger intervalterminates; waiting for the new acquisition process to terminate duringa late trigger interval that starts at the time when the expectedtrigger interval terminated; and if the late trigger interval expiresand the new acquisition process has not terminated, aborting the newacquisition process and forwarding whatever rasterization data isavailable.
 2. A method for maintaining rasterizer functionalityaccording to claim 1 wherein the aborting step comprises the steps of:determining from the trigger circuit at the expiration of the latetrigger interval whether a trigger has been received; and if the triggerhas been received, allowing the new acquisition to proceed tocompletion; or if the trigger has not been received, aborting the newacquisition process.
 3. A method for maintaining rasterizationfunctionality according to claim 1 wherein the late trigger interval ofthe waiting step is non-uniform.
 4. A method for maintainingrasterization functionality according to claim 2 wherein the latetrigger interval of the waiting step is non-uniform.
 5. A method formaintaining rasterization functionality according to claim 3 wherein thenon-uniform late trigger interval is made to vary within a predeterminedrange.
 6. A method for maintaining rasterization functionality accordingto claim 4 wherein the non-uniform late trigger interval is made to varywithin a predetermined range.
 7. A method for maintaining rasterizationfunctionality according to claim 3 wherein the non-uniform late triggerinterval is made to vary in a random manner within a predeterminedrange.
 8. A method for maintaining rasterization functionality accordingto claim 4 wherein the non-uniform late trigger interval is made to varyin a random manner within a predetermined range.